Method and apparatus for adaptive tuning of wireless power transfer

ABSTRACT

Exemplary embodiments are directed to wireless power transfer. A transmit antenna generates an electromagnetic field at a resonant frequency of to create a coupling-mode region within a near field of the transmit antenna. A receive antenna receives the resonant frequency when it is within the coupling-mode region and resonates substantially near the resonant frequency. One, or both, of the transmit and receive antennas are tunable antennas that can be adaptively tuned. The adaptive tuning is accomplished by detecting a mismatch at the tunable antenna and generating a mismatch signal responsive to a voltage standing wave ratio at the tunable antenna. A resonance characteristic of the tunable antenna can be modified by adjusting a capacitance of a variable capacitor network connected to the tunable antenna.

CLAIM OF PRIORITY UNDER 35 U.S.C. §119

This application claims priority under 35 U.S.C. §119(e) to:

-   -   U.S. Provisional Patent Application 61/060,735 entitled “REVERSE         LINK SIGNALING VIA RECEIVE ANTENNA IMPEDANCE MODULATION” filed         on Jun. 11, 2008;     -   U.S. Provisional Patent Application 61/060,738 entitled         “SIGNALING CHARGING IN WIRELESS POWER ENVIRONMENT” filed on Jun.         11, 2008;     -   U.S. Provisional Patent Application 61/053,008 entitled         “ADAPTIVE TUNING MECHANISM FOR WIRELESS POWER TRANSFER” filed on         May 13, 2008;     -   U.S. Provisional Patent Application 61/053,010 entitled         “EFFICIENT POWER MANAGEMENT SCHEME FOR WIRELESS POWER CHARGING         SYSTEMS” filed on May 13, 2008;     -   U.S. Provisional Patent Application 61/060,741 entitled         “TRANSMIT POWER CONTROL FOR A WIRELESS CHARGING SYSTEM” filed on         Jun. 11, 2008;     -   U.S. Provisional Patent Application 61/053,000 entitled         “REPEATERS FOR ENHANCEMENT OF WIRELESS POWER TRANSFER” filed on         May 13, 2008;     -   U.S. Provisional Patent Application 61/053,004 entitled         “WIRELESS POWER TRANSFER FOR APPLIANCES AND EQUIPMENTS” filed on         May 13, 2008;     -   U.S. Provisional Patent Application 61/081,332 entitled         “WIRELESS POWER TRANSFER USING NEGATIVE RESISTANCE” filed on         Jul. 16, 2008;     -   U.S. Provisional Patent Application 61/053,012 entitled         “EMBEDDED RECEIVE ANTENNA FOR WIRELESS POWER TRANSFER” filed on         May 13, 2008; and     -   U.S. Provisional Patent Application 61/053,015 entitled “PLANAR         LARGE AREA WIRELESS CHARGING SYSTEM” filed on May 13, 2008.

REFERENCE TO CO-PENDING APPLICATIONS FOR PATENT

This application is also related to the following applications, which are assigned to the assignee hereof and filed on even date herewith, the disclosures of which are incorporated herein in their entirety by reference.

-   -   U.S. Patent Application (Attorney Docket 080793) entitled         “REPEATERS FOR ENHANCEMENT OF WIRELESS POWER TRANSFER.”     -   U.S. Patent Application (Attorney Docket 080253) entitled         “WIRELESS POWER TRANSFER FOR APPLIANCES AND EQUIPMENTS.”     -   U.S. Patent Application (Attorney Docket 081046) entitled         “METHOD AND APPARATUS WITH NEGATIVE RESISTANCE IN WIRELESS POWER         TRANSFERS.”

BACKGROUND

Typically, each battery powered device such as a wireless electronic device requires its own charger and power source, which is usually an alternating current (AC) power outlet. Such a wired configuration becomes unwieldy when many devices need charging.

Approaches are being developed that use over-the-air or wireless power transmission between a transmitter and a receiver coupled to the electronic device to be charged. Such approaches generally fall into two categories. One is based on the coupling of plane wave radiation (also called far-field radiation) between a transmit antenna and a receive antenna on the device to be charged. The receive antenna collects the radiated power and rectifies it for charging the battery. Antennas are generally of resonant length in order to improve the coupling efficiency. This approach suffers from the fact that the power coupling falls off quickly with distance between the antennas. So charging over reasonable distances (e.g., less than 1 to 2 meters) becomes difficult. Additionally, since the transmitting system radiates plane waves, unintentional radiation can interfere with other systems if not properly controlled through filtering.

Other approaches to wireless energy transmission techniques are based on inductive coupling between a transmit antenna embedded, for example, in a “charging” mat or surface and a receive antenna (plus a rectifying circuit) embedded in the host electronic device to be charged. This approach has the disadvantage that the spacing between transmit and receive antennas must be very close (e.g., within thousandths of meters). Though this approach does have the capability to simultaneously charge multiple devices in the same area, this area is typically very small and requires the user to accurately locate the devices to a specific area. Therefore, there is a need to provide a wireless charging arrangement that accommodates flexible placement and orientation of transmit and receive antennas.

With wireless power transmission there is a need for systems and methods for adjusting the operating characteristics of the antennas to adapt to different circumstances and optimize power transfer characteristics.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a simplified block diagram of a wireless power transfer system.

FIG. 2 shows a simplified schematic diagram of a wireless power transfer system.

FIG. 3 shows a schematic diagram of a loop antenna for use in exemplary embodiments of the present invention.

FIG. 4 shows simulation results indicating coupling strength between transmit and receive antennas.

FIGS. 5A and 5B show layouts of loop antennas for transmit and receive antennas according to exemplary embodiments of the present invention.

FIG. 6 shows simulation results indicating coupling strength between transmit and receive antennas relative to various circumference sizes for the square and circular transmit antennas illustrated in FIGS. 5A and 5B.

FIG. 7 shows simulation results indicating coupling strength between transmit and receive antennas relative to various surface areas for the square and circular transmit antennas illustrated in FIGS. 5A and 5B.

FIG. 8 shows various placement points for a receive antenna relative to a transmit antenna to illustrate coupling strengths in coplanar and coaxial placements.

FIG. 9 shows simulation results indicating coupling strength for coaxial placement at various distances between the transmit and receive antennas.

FIG. 10 is a simplified block diagram of a transmitter, in accordance with an exemplary embodiment of the present invention.

FIG. 11 is a simplified block diagram of a receiver, in accordance with an exemplary embodiment of the present invention.

FIG. 12 shows a simplified schematic of a portion of transmit circuitry for carrying out messaging between a transmitter and a receiver.

FIGS. 13A-13C shows a simplified schematic of a portion of receive circuitry in various states to illustrate messaging between a receiver and a transmitter.

FIGS. 14A-14C shows a simplified schematic of a portion of alternative receive circuitry in various states to illustrate messaging between a receiver and a transmitter.

FIGS. 15A-15C are timing diagrams illustrating a messaging protocol for communication between a transmitter and a receiver.

FIGS. 16A-16D are simplified block diagrams illustrating a beacon power mode for transmitting power between a transmitter and a receiver.

FIG. 17A illustrates a large transmit antenna with a smaller repeater antenna disposed coplanar with, and coaxial with, the transmit antenna.

FIG. 17B illustrates a transmit antenna with a larger repeater antenna with a coaxial placement relative to the transmit antenna.

FIG. 18A illustrates a large transmit antenna with a three different smaller repeater antennas disposed coplanar with, and within a perimeter of, the transmit antenna.

FIG. 18B illustrates a large transmit antenna with smaller repeater antennas with offset coaxial placements and offset coplanar placements relative to the transmit antenna.

FIG. 19 shows simulation results indicating coupling strength between a transmit antenna, a repeater antenna and a receive antenna.

FIG. 20A shows simulation results indicating coupling strength between a transmit antenna and receive antenna with no repeater antennas.

FIG. 20B shows simulation results indicating coupling strength between a transmit antenna and receive antenna with a repeater antenna.

FIG. 21A-21C are simplified block diagrams of adaptive tuning circuits for an antenna using a T-network, an L-network, and a Pi-network, respectively.

FIG. 22 is a simplified block diagram of an adaptive tuning circuit for a transmit antenna based on power consumption at the transmit antenna.

FIGS. 23A and 23B are simplified circuit diagrams illustrating exemplary embodiments of variable capacitor networks.

FIGS. 24A and 24B show simulation results for near field coupled transmit and receive antennas before and after adaptive tuning, respectively.

DETAILED DESCRIPTION

The word “exemplary” is used herein to mean “serving as an example, instance, or illustration.” Any embodiment described herein as “exemplary” is not necessarily to be construed as preferred or advantageous over other embodiments.

The detailed description set forth below in connection with the appended drawings is intended as a description of exemplary embodiments of the present invention and is not intended to represent the only embodiments in which the present invention can be practiced. The term “exemplary” used throughout this description means “serving as an example, instance, or illustration,” and should not necessarily be construed as preferred or advantageous over other exemplary embodiments. The detailed description includes specific details for the purpose of providing a thorough understanding of the exemplary embodiments of the invention. It will be apparent to those skilled in the art that the exemplary embodiments of the invention may be practiced without these specific details. In some instances, well-known structures and devices are shown in block diagram form in order to avoid obscuring the novelty of the exemplary embodiments presented herein.

The words “wireless power” is used herein to mean any form of energy associated with electric fields, magnetic fields, electromagnetic fields, or otherwise that is transmitted between from a transmitter to a receiver without the use of physical electromagnetic conductors.

FIG. 1 illustrates wireless transmission or charging system 100, in accordance with various exemplary embodiments of the present invention. Input power 102 is provided to a transmitter 104 for generating a radiated field 106 for providing energy transfer. A receiver 108 couples to the radiated field 106 and generates an output power 110 for storing or consumption by a device (not shown) coupled to the output power 110. Both the transmitter 104 and the receiver 108 are separated by a distance 112. In one exemplary embodiment, transmitter 104 and receiver 108 are configured according to a mutual resonant relationship and when the resonant frequency of receiver 108 and the resonant frequency of transmitter 104 are exactly identical, transmission losses between the transmitter 104 and the receiver 108 are minimal when the receiver 108 is located in the “near-field” of the radiated field 106.

Transmitter 104 further includes a transmit antenna 114 for providing a means for energy transmission and receiver 108 further includes a receive antenna 118 for providing a means for energy reception. The transmit and receive antennas are sized according to applications and devices to be associated therewith. As stated, an efficient energy transfer occurs by coupling a large portion of the energy in the near-field of the transmitting antenna to a receiving antenna rather than propagating most of the energy in an electromagnetic wave to the far field. When in this near-field a coupling mode may be developed between the transmit antenna 114 and the receive antenna 118. The area around the antennas 114 and 118 where this near-field coupling may occur is referred to herein as a coupling-mode region.

FIG. 2 shows a simplified schematic diagram of a wireless power transfer system. The transmitter 104 includes an oscillator 122, a power amplifier 124 and a filter and matching circuit 126. The oscillator is configured to generate at a desired frequency, which may be adjusted in response to adjustment signal 123. The oscillator signal may be amplified by the power amplifier 124 with an amplification amount responsive to control signal 125. The filter and matching circuit 126 may be included to filter out harmonics or other unwanted frequencies and match the impedance of the transmitter 104 to the transmit antenna 114.

The receiver may include a matching circuit 132 and a rectifier and switching circuit to generate a DC power output to charge a battery 136 as shown in FIG. 2 or power a device coupled to the receiver (not shown). The matching circuit 132 may be included to match the impedance of the receiver 108 to the receive antenna 118.

As illustrated in FIG. 3, antennas used in exemplary embodiments may be configured as a “loop” antenna 150, which may also be referred to herein as a “magnetic” antenna. Loop antennas may be configured to include an air core or a physical core such as a ferrite core. Air core loop antennas may be more tolerable to extraneous physical devices placed in the vicinity of the core. Furthermore, an air core loop antenna allows the placement of other components within the core area. In addition, an air core loop may more readily enable placement of the receive antenna 118 (FIG. 2) within a plane of the transmit antenna 114 (FIG. 2) where the coupled-mode region of the transmit antenna 114 (FIG. 2) may be more powerful.

As stated, efficient transfer of energy between the transmitter 104 and receiver 108 occurs during matched or nearly matched resonance between the transmitter 104 and the receiver 108. However, even when resonance between the transmitter 104 and receiver 108 are not matched, energy may be transferred at a lower efficiency. Transfer of energy occurs by coupling energy from the near-field of the transmitting antenna to the receiving antenna residing in the neighborhood where this near-field is established rather than propagating the energy from the transmitting antenna into free space.

The resonant frequency of the loop or magnetic antennas is based on the inductance and capacitance. Inductance in a loop antenna is generally simply the inductance created by the loop, whereas, capacitance is generally added to the loop antenna's inductance to create a resonant structure at a desired resonant frequency. As a non-limiting example, capacitor 152 and capacitor 154 may be added to the antenna to create a resonant circuit that generates resonant signal 156. Accordingly, for larger diameter loop antennas, the size of capacitance needed to induce resonance decreases as the diameter or inductance of the loop increases. Furthermore, as the diameter of the loop or magnetic antenna increases, the efficient energy transfer area of the near-field increases. Of course, other resonant circuits are possible. As another non-limiting example, a capacitor may be placed in parallel between the two terminals of the loop antenna. In addition, those of ordinary skill in the art will recognize that for transmit antennas the resonant signal 156 may be an input to the loop antenna 150.

Exemplary embodiments of the invention include coupling power between two antennas that are in the near-fields of each other. As stated, the near-field is an area around the antenna in which electromagnetic fields exist but may not propagate or radiate away from the antenna. They are typically confined to a volume that is near the physical volume of the antenna. In the exemplary embodiments of the invention, magnetic type antennas such as single and multi-turn loop antennas are used for both transmit (Tx) and receive (Rx) antenna systems since magnetic near-field amplitudes tend to be higher for magnetic type antennas in comparison to the electric near-fields of an electric-type antenna (e.g., a small dipole). This allows for potentially higher coupling between the pair. Furthermore, “electric” antennas (e.g., dipoles and monopoles) or a combination of magnetic and electric antennas is also contemplated.

The Tx antenna can be operated at a frequency that is low enough and with an antenna size that is large enough to achieve good coupling (e.g., >−4 dB) to a small Rx antenna at significantly larger distances than allowed by far field and inductive approaches mentioned earlier. If the Tx antenna is sized correctly, high coupling levels (e.g., −2 to −4 dB) can be achieved when the Rx antenna on a host device is placed within a coupling-mode region (i.e., in the near-field) of the driven Tx loop antenna.

FIG. 4 shows simulation results indicating coupling strength between transmit and receive antennas. Curves 170 and 172 indicate a measure of acceptance of power by the transmit and receive antennas, respectively. In other words, with a large negative number there is a very close impedance match and most of the power is accepted and, as a result, radiated by the transmit antenna. Conversely, a small negative number indicates that much of the power is reflected back from the antenna because there is not a close impedance match at the given frequency. In FIG. 4, the transmit antenna and the receive antenna are tuned to have a resonant frequency of about 13.56 MHz.

Curve 170 illustrates the amount of power transmitted from the transmit antenna at various frequencies. Thus, at points 1 a and 3 a, corresponding to about 13.528 MHz and 13.593 MHz, much of the power is reflected and not transmitted out of the transmit antenna. However, at point 2 a, corresponding to about 13.56 MHz, it can be seen that a large amount of the power is accepted and transmitted out of the antenna.

Similarly, curve 172 illustrates the amount of power received by the receive antenna at various frequencies. Thus, at points 1 b and 3 b, corresponding to about 13.528 MHz and 13.593 MHz, much of the power is reflected and not conveyed through the receive antenna and into the receiver. However, at point 2 b corresponding to about 13.56 MHz, it can be seen that a large amount of the power is accepted by the receive antenna and conveyed into the receiver.

Curve 174 indicates the amount of power received at the receiver after being sent from the transmitter through the transmit antenna, received through the receive antenna and conveyed to the receiver. Thus, at points 1 c and 3 c, corresponding to about 13.528 MHz and 13.593 MHz, much of the power sent out of the transmitter is not available at the receiver because (1) the transmit antenna rejects much of the power sent to it from the transmitter and (2) the coupling between the transmit antenna and the receive antenna is less efficient as the frequencies move away from the resonant frequency. However, at point 2 c corresponding to about 13.56 MHz, it can be seen that a large amount of the power sent from the transmitter is available at the receiver, indicating a high degree of coupling between the transmit antenna and the receive antenna.

FIGS. 5A and 5B show layouts of loop antennas for transmit and receive antennas according to exemplary embodiments of the present invention. Loop antennas may be configured in a number of different ways, with single loops or multiple loops at wide variety of sizes. In addition, the loops may be a number of different shapes, such as, for example only, circular, elliptical, square, and rectangular. FIG. 5A illustrates a large square loop transmit antenna 114S and a small square loop receive antenna 118 placed in the same plane as the transmit antenna 114S and near the center of the transmit antenna 114S. FIG. 5B illustrates a large circular loop transmit antenna 114C and a small square loop receive antenna 118′ placed in the same plane as the transmit antenna 114C and near the center of the transmit antenna 114C. The square loop transmit antenna 114S has side lengths of “a” while the circular loop transmit antenna 114C has a diameter of “Φ.” For a square loop, it can be shown that there is an equivalent circular loop whose diameter may be defined as: Φ_(eq)=4a/π.

FIG. 6 shows simulation results indicating coupling strength between transmit and receive antennas relative to various circumferences for the square and circular transmit antennas illustrated in FIGS. 4A and 4B. Thus, curve 180 shows coupling strength between the circular loop transmit antennas 114C and the receive antenna 118 at various circumference sizes for the circular loop transmit antenna 114C. Similarly, curve 182 shows coupling strength between the square loop transmit antennas 114S and the receive antenna 118′ at various equivalent circumference sizes for the transmit loop transmit antenna 114S.

FIG. 7 shows simulation results indicating coupling strength between transmit and receive antennas relative to various surface areas for the square and circular transmit antennas illustrated in FIGS. 5A and 5B. Thus, curve 190 shows coupling strength between the circular loop transmit antennas 114C and the receive antenna 118 at various surface areas for the circular loop transmit antenna 114C. Similarly, curve 192 shows coupling strength between the square loop transmit antennas 114S and the receive antenna 118′ at various surface areas for the transmit loop transmit antenna 114S.

FIG. 8 shows various placement points for a receive antenna relative to a transmit antenna to illustrate coupling strengths in coplanar and coaxial placements. “Coplanar,” as used herein, means that the transmit antenna and receive antenna have planes that are substantially aligned (i.e., have surface normals pointing in substantially the same direction) and with no distance (or a small distance) between the planes of the transmit antenna and the receive antenna. “Coaxial,” as used herein, means that the transmit antenna and receive antenna have planes that are substantially aligned (i.e., have surface normals pointing in substantially the same direction) and the distance between the two planes is not trivial and furthermore, the surface normal of the transmit antenna and the receive antenna lie substantially along the same vector, or the two normals are in echelon.

As examples, points p1, p2, p3, and p7 are all coplanar placement points for a receive antenna relative to a transmit antenna. As another example, point p5 and p6 are coaxial placement points for a receive antenna relative to a transmit antenna. The table below shows coupling strength (S21) and coupling efficiency (expressed as a percentage of power transmitted from the transmit antenna that reached the receive antenna) at the various placement points (p1-p7) illustrated in FIG. 8.

TABLE 1 Efficiency (TX DC power in to Distance from S21 efficiency RX DC power Position plane (cm) (%) out) p1 0 46.8 28 p2 0 55.0 36 p3 0 57.5 35 p4 2.5 49.0 30 p5 17.5 24.5 15 p6 17.5 0.3 0.2 p7 0 5.9 3.4

As can be seen, the coplanar placement points p1, p2, and p3, all show relatively high coupling efficiencies. Placement point p7 is also a coplanar placement point, but is outside of the transmit loop antenna. While placement point p7 does not have a high coupling efficiency, it is clear that there is some coupling and the coupling-mode region extends beyond the perimeter of the transmit loop antenna.

Placement point p5 is coaxial with the transmit antenna and shows substantial coupling efficiency. The coupling efficiency for placement point p5 is not as high as the coupling efficiencies for the coplanar placement points. However, the coupling efficiency for placement point p5 is high enough that substantial power can be conveyed between the transmit antenna and a receive antenna in a coaxial placement.

Placement point p4 is within the circumference of the transmit antenna but at a slight distance above the plane of the transmit antenna in a position that may be referred to as an offset coaxial placement (i.e., with surface normals in substantially the same direction but at different locations) or offset coplanar (i.e., with surface normals in substantially the same direction but with planes that are offset relative to each other). From the table it can be seen that with an offset distance of 2.5 cm, placement point p4 still has relatively good coupling efficiency.

Placement point p6 illustrates a placement point outside the circumference of the transmit antenna and at a substantial distance above the plane of the transmit antenna. As can be seen from the table, placement point p7 shows little coupling efficiency between the transmit and receive antennas.

FIG. 9 shows simulation results indicating coupling strength for coaxial placement at various distances between the transmit and receive antennas. The simulations for FIG. 9 are for square transmit and receive antennas in a coaxial placement, both with sides of about 1.2 meters and at a transmit frequency of 10 MHz. It can be seen that the coupling strength remains quite high and uniform at distances of less than about 0.5 meters.

FIG. 10 is a simplified block diagram of a transmitter, in accordance with an exemplary embodiment of the present invention. A transmitter 200 includes transmit circuitry 202 and a transmit antenna 204. Generally, transmit circuitry 202 provides RF power to the transmit antenna 204 by providing an oscillating signal resulting in generation of near-field energy about the transmit antenna 204. By way of example, transmitter 200 may operate at the 13.56 MHz ISM band.

Exemplary transmit circuitry 202 includes a fixed impedance matching circuit 206 for matching the impedance of the transmit circuitry 202 (e.g., 50 ohms) to the transmit antenna 204 and a low pass filter (LPF) 208 configured to reduce harmonic emissions to levels to prevent self-jamming of devices coupled to receivers 108 (FIG. 1). Other embodiments may include different filter topologies, including but not limited to, notch filters that attenuate specific frequencies while passing others and may include an adaptive impedance match, that can be varied based on measurable transmit metrics, such as output power to the antenna or DC current draw by the power amplifier. Transmit circuitry 202 further includes a power amplifier 210 configured to drive an RF signal as determined by an oscillator 212. The transmit circuitry may be comprised of discrete devices or circuits, or alternately, may be comprised of an integrated assembly. An exemplary RF power output from transmit antenna 204 may be on the order of 2.5 Watts.

Transmit circuitry 202 further includes a processor 214 for enabling the oscillator 212 during transmit phases (or duty cycles) for specific receivers, for adjusting the frequency of the oscillator, and for adjusting the output power level for implementing a communication protocol for interacting with neighboring devices through their attached receivers.

The transmit circuitry 202 may further include a load sensing circuit 216 for detecting the presence or absence of active receivers in the vicinity of the near-field generated by transmit antenna 204. By way of example, a load sensing circuit 216 monitors the current flowing to the power amplifier 210, which is affected by the presence or absence of active receivers in the vicinity of the near-field generated by transmit antenna 204. Detection of changes to the loading on the power amplifier 210 are monitored by processor 214 for use in determining whether to enable the oscillator 212 for transmitting energy to communicate with an active receiver.

Transmit antenna 204 may be implemented as an antenna strip with the thickness, width and metal type selected to keep resistive losses low. In a conventional implementation, the transmit antenna 204 can generally be configured for association with a larger structure such as a table, mat, lamp or other less portable configuration. Accordingly, the transmit antenna 204 generally will not need “turns” in order to be of a practical dimension. An exemplary implementation of a transmit antenna 204 may be “electrically small” (i.e., fraction of the wavelength) and tuned to resonate at lower usable frequencies by using capacitors to define the resonant frequency. In an exemplary application where the transmit antenna 204 may be larger in diameter, or length of side if a square loop, (e.g., 0.50 meters) relative to the receive antenna, the transmit antenna 204 will not necessarily need a large number of turns to obtain a reasonable capacitance.

FIG. 11 is a block diagram of a receiver, in accordance with an embodiment of the present invention. A receiver 300 includes receive circuitry 302 and a receive antenna 304. Receiver 300 further couples to device 350 for providing received power thereto. It should be noted that receiver 300 is illustrated as being external to device 350 but may be integrated into device 350. Generally, energy is propagated wirelessly to receive antenna 304 and then coupled through receive circuitry 302 to device 350.

Receive antenna 304 is tuned to resonate at the same frequency, or near the same frequency, as transmit antenna 204 (FIG. 10). Receive antenna 304 may be similarly dimensioned with transmit antenna 204 or may be differently sized based upon the dimensions of an associated device 350. By way of example, device 350 may be a portable electronic device having diametric or length dimension smaller that the diameter of length of transmit antenna 204. In such an example, receive antenna 304 may be implemented as a multi-turn antenna in order to reduce the capacitance value of a tuning capacitor (not shown) and increase the receive antenna's impedance. By way of example, receive antenna 304 may be placed around the substantial circumference of device 350 in order to maximize the antenna diameter and reduce the number of loop turns (i.e., windings) of the receive antenna and the inter-winding capacitance.

Receive circuitry 302 provides an impedance match to the receive antenna 304. Receive circuitry 302 includes power conversion circuitry 306 for converting a received RF energy source into charging power for use by device 350. Power conversion circuitry 306 includes an RF-to-DC converter 308 and may also in include a DC-to-DC converter 310. RF-to-DC converter 308 rectifies the RF energy signal received at receive antenna 304 into a non-alternating power while DC-to-DC converter 310 converts the rectified RF energy signal into an energy potential (e.g., voltage) that is compatible with device 350. Various RF-to-DC converters are contemplated including partial and full rectifiers, regulators, bridges, doublers, as well as linear and switching converters.

Receive circuitry 302 may further include switching circuitry 312 for connecting receive antenna 304 to the power conversion circuitry 306 or alternatively for disconnecting the power conversion circuitry 306. Disconnecting receive antenna 304 from power conversion circuitry 306 not only suspends charging of device 350, but also changes the “load” as “seen” by the transmitter 200 (FIG. 2) as is explained more fully below. As disclosed above, transmitter 200 includes load sensing circuit 216 which detects fluctuations in the bias current provided to transmitter power amplifier 210. Accordingly, transmitter 200 has a mechanism for determining when receivers are present in the transmitter's near-field.

When multiple receivers 300 are present in a transmitter's near-field, it may be desirable to time-multiplex the loading and unloading of one or more receivers to enable other receivers to more efficiently couple to the transmitter. A receiver may also be cloaked in order to eliminate coupling to other nearby receivers or to reduce loading on nearby transmitters. This “unloading” of a receiver is also known herein as a “cloaking.” Furthermore, this switching between unloading and loading controlled by receiver 300 and detected by transmitter 200 provides a communication mechanism from receiver 300 to transmitter 200 as is explained more fully below. Additionally, a protocol can be associated with the switching which enables the sending of a message from receiver 300 to transmitter 200. By way of example, a switching speed may be on the order of 100 μsec.

In an exemplary embodiment, communication between the transmitter and the receiver refers to a Device Sensing and Charging Control Mechanism, rather than conventional two-way communication. In other words, the transmitter uses on/off keying of the transmitted signal to adjust whether energy is available in the near-filed. The receivers interpret these changes in energy as a message from the transmitter. From the receiver side, the receiver uses tuning and de-tuning of the receive antenna to adjust how much power is being accepted from the near-field. The transmitter can detect this difference in power used from the near field and interpret these changes as a message from the receiver.

Receive circuitry 302 may further include signaling detector and beacon circuitry 314 used to identify received energy fluctuations, which may correspond to informational signaling from the transmitter to the receiver. Furthermore, signaling and beacon circuitry 314 may also be used to detect the transmission of a reduced RF signal energy (i.e., a beacon signal) and to rectify the reduced RF signal energy into a nominal power for awakening either un-powered or power-depleted circuits within receive circuitry 302 in order to configure receive circuitry 302 for wireless charging.

Receive circuitry 302 further includes processor 316 for coordinating the processes of receiver 300 described herein including the control of switching circuitry 312 described herein. Cloaking of receiver 300 may also occur upon the occurrence of other events including detection of an external wired charging source (e.g., wall/USB power) providing charging power to device 350. Processor 316, in addition to controlling the cloaking of the receiver, may also monitor beacon circuitry 314 to determine a beacon state and extract messages sent from the transmitter. Processor 316 may also adjust DC-to-DC converter 310 for improved performance.

FIG. 12 shows a simplified schematic of a portion of transmit circuitry for carrying out messaging between a transmitter and a receiver. In some exemplary embodiments of the present invention, a means for communication may be enabled between the transmitter and the receiver. In FIG. 12 a power amplifier 210 drives the transmit antenna 204 to generate the radiated field. The power amplifier is driven by a carrier signal 220 that is oscillating at a desired frequency for the transmit antenna 204. A transmit modulation signal 224 is used to control the output of the power amplifier 210.

The transmit circuitry can send signals to receivers by using an ON/OFF keying process on the power amplifier 210. In other words, when the transmit modulation signal 224 is asserted, the power amplifier 210 will drive the frequency of the carrier signal 220 out on the transmit antenna 204. When the transmit modulation signal 224 is negated, the power amplifier will not drive out any frequency on the transmit antenna 204.

The transmit circuitry of FIG. 12 also includes a load sensing circuit 216 that supplies power to the power amplifier 210 and generates a receive signal 235 output. In the load sensing circuit 216 a voltage drop across resistor R_(s) develops between the power in signal 226 and the power supply 228 to the power amplifier 210. Any change in the power consumed by the power amplifier 210 will cause a change in the voltage drop that will be amplified by differential amplifier 230. When the transmit antenna is in coupled mode with a receive antenna in a receiver (not shown in FIG. 12) the amount of current drawn by the power amplifier 210 will change. In other words, if no coupled mode resonance exist for the transmit antenna 210, the power required to drive the radiated field will be first amount. If a coupled mode resonance exists, the amount of power consumed by the power amplifier 210 will go up because much of the power is being coupled into the receive antenna. Thus, the receive signal 235 can indicate the presence of a receive antenna coupled to the transmit antenna 235 and can also detect signals sent from the receive antenna, as explained below. Additionally, a change in receiver current draw will be observable in the transmitter's power amplifier current draw, and this change can be used to detect signals from the receive antennas, as explained below.

FIGS. 13A-13C shows a simplified schematic of a portion of receive circuitry in various states to illustrate messaging between a receiver and a transmitter. All of FIGS. 13A-13C show the same circuit elements with the difference being state of the various switches. A receive antenna 304 includes a characteristic inductance L1, which drives node 350. Node 350 is selectively coupled to ground through switch S1A. Node 350 is also selectively coupled to diode D1 and rectifier 318 through switch S1B. The rectifier 318 supplies a DC power signal 322 to a receive device (not shown) to power the receive device, charge a battery, or a combination thereof. The diode D1 is coupled to a transmit signal 320 which is filtered to remove harmonics and unwanted frequencies with capacitor C3 and resistor R1. Thus the combination of D1, C3, and R1 can generate a signal on the transmit signal 320 that mimics the transmit modulation generated by the transmit modulation signal 224 discussed above with reference to the transmitter in FIG. 12.

Exemplary embodiments of the invention includes modulation of the receive device's current draw and modulation of the receive antenna's impedance to accomplish reverse link signaling. With reference to both FIG. 13A and FIG. 12, as the power draw of the receive device changes, the load sensing circuit 216 detects the resulting power changes on the transmit antenna and from these changes can generate the receive signal 235.

In the embodiments of FIGS. 13A-13C, the current draw through the transmitter can be changed by modifying the state of switches S1A and S2A. In FIG. 13A, switch S1A and switch S2A are both open creating a “DC open state” and essentially removing the load from the transmit antenna 204. This reduces the current seen by the transmitter.

In FIG. 13B, switch S1A is closed and switch S2A is open creating a “DC short state” for the receive antenna 304. Thus the state in FIG. 13B can be used to increase the current seen in the transmitter.

In FIG. 13C, switch S1A is open and switch S2A is closed creating a normal receive mode (also referred to herein as a “DC operating state”) wherein power can be supplied by the DC out signal 322 and a transmit signal 320 can be detected. In the state shown in FIG. 13C the receiver receives a normal amount of power, thus consuming more or less power from the transmit antenna than the DC open state or the DC short state.

Reverse link signaling may be accomplished by switching between the DC operating state (FIG. 13C) and the DC short state (FIG. 13B). Reverse link signaling also may be accomplished by switching between the DC operating state (FIG. 13C) and the DC open state (FIG. 13A).

FIGS. 14A-14C shows a simplified schematic of a portion of alternative receive circuitry in various states to illustrate messaging between a receiver and a transmitter.

All of FIGS. 14A-14C show the same circuit elements with the difference being state of the various switches. A receive antenna 304 includes a characteristic inductance L1, which drives node 350. Node 350 is selectively coupled to ground through capacitor C1 and switch S1B. Node 350 is also AC coupled to diode D1 and rectifier 318 through capacitor C2. The diode D1 is coupled to a transmit signal 320 which is filtered to remove harmonics and unwanted frequencies with capacitor C3 and resistor R1. Thus the combination of D1, C3, and R1 can generate a signal on the transmit signal 320 that mimics the transmit modulation generated by the transmit modulation signal 224 discussed above with reference to the transmitter in FIG. 12.

The rectifier 318 is connected to switch S2B, which is connected in series with resistor R2 and ground. The rectifier 318 also is connected to switch S3B. The other side of switch S3B supplies a DC power signal 322 to a receive device (not shown) to power the receive device, charge a battery, or a combination thereof.

In FIGS. 13A-13C the DC impedance of the receive antenna 304 is changed by selectively coupling the receive antenna to ground through switch S1B. In contrast, in the embodiments of FIGS. 14A-14C, the impedance of the antenna can be modified to generate the reverse link signaling by modifying the state of switches S1B, S2B, and S3B to change the AC impedance of the receive antenna 304. In FIGS. 14A-14C the resonant frequency of the receive antenna 304 may be tuned with capacitor C2. Thus, the AC impedance of the receive antenna 304 may be changed by selectively coupling the receive antenna 304 through capacitor C1 using switch S1B, essentially changing the resonance circuit to a different frequency that will be outside of a range that will optimally couple with the transmit antenna. If the resonance frequency of the receive antenna 304 is near the resonant frequency of the transmit antenna, and the receive antenna 304 is in the near-field of the transmit antenna, a coupling mode may develop wherein the receiver can draw significant power from the radiated field 106.

In FIG. 14A, switch S1B is closed, which de-tunes the antenna and creates an “AC cloaking state,” essentially “cloaking” the receive antenna 304 from detection by the transmit antenna 204 because the receive antenna does not resonate at the transmit antenna's frequency. Since the receive antenna will not be in a coupled mode, the state of switches S2B and S3B are not particularly important to the present discussion.

In FIG. 14B, switch S1B is open, switch S2B is closed, and switch S3B is open, creating a “tuned dummy-load state” for the receive antenna 304. Because switch S1B is open, capacitor C1 does not contribute to the resonance circuit and the receive antenna 304 in combination with capacitor C2 will be in a resonance frequency that may match with the resonant frequency of the transmit antenna. The combination of switch S3B open and switch S2B closed creates a relatively high current dummy load for the rectifier, which will draw more power through the receive antenna 304, which can be sensed by the transmit antenna. In addition, the transmit signal 320 can be detected since the receive antenna is in a state to receive power from the transmit antenna.

In FIG. 14C, switch S1B is open, switch S2B is open, and switch S3B is closed, creating a “tuned operating state” for the receive antenna 304. Because switch S1B is open, capacitor C1 does not contribute to the resonance circuit and the receive antenna 304 in combination with capacitor C2 will be in a resonance frequency that may match with the resonant frequency of the transmit antenna. The combination of switch S2B open and switch S3B closed creates a normal operating state wherein power can be supplied by the DC out signal 322 and a transmit signal 320 can be detected.

Reverse link signaling may be accomplished by switching between the tuned operating state (FIG. 14C) and the AC cloaking state (FIG. 14A). Reverse link signaling also may be accomplished by switching between the tuned dummy-load state (FIG. 14B) and the AC cloaking state (FIG. 14A). Reverse link signaling also may be accomplished by switching between the tuned operating state (FIG. 14C) and the tuned dummy-load state (FIG. 14B) because there will be a difference in the amount of power consumed by the receiver, which can be detected by the load sensing circuit in the transmitter.

Of course, those of ordinary skill in the art will recognize that other combinations of switches S1B, S2B, and S3B may be used to create cloaking, generate reverse link signaling and supplying power to the receive device. In addition, the switches S1A and S1B may be added to the circuits of FIGS. 14A-14C to create other possible combinations for cloaking, reverse link signaling, and supplying power to the receive device.

Thus, when in a coupled mode signals may be sent from the transmitter to the receiver, as discussed above with reference to FIG. 12. In addition, when in a coupled mode signals may be sent from the receiver to the transmitter, as discussed above with reference to FIGS. 13A-13C and 14A-14C.

FIGS. 15A-15C are timing diagrams illustrating a messaging protocol for communication between a transmitter and a receiver using the signaling techniques discussed above. In one exemplary approach, signals from the transmitter to the receiver are referred to herein as a “forward link” and use a simple AM modulation between normal oscillation and no oscillation. Other modulation techniques are also contemplated. As a non-limiting example, a signal present may be interpreted as a 1 and no signal present may be interpreted as a 0.

Reverse link signaling is provided by modulation of power drawn by the receive device, which can be detected by the load sensing circuit in the transmitter. As a non-limiting example, higher power states may be interpreted as a 1 and lower power states may be interpreted as a 0. It should be noted that the transmitter must be on for the receiver to be able to perform the reverse link signaling. In addition, the receiver should not perform reverse link signaling during forward link signaling. Furthermore, if two receive devices attempt to perform reverse link signaling at the same time a collision may occur, which will make it difficult, if not impossible for the transmitter to decode a proper reverse link signal.

In the exemplary embodiment described herein, signaling is similar to a Universal Asynchronous Receive Transmit (UART) serial communication protocol with a start bit, a data byte, a parity bit and a stop bit. Of course, any serial communication protocol may be suitable for carrying the exemplary embodiment of the present invention described herein. For simplicity of description, and not as a limitation, the messaging protocol will be described such that the period for communicating each byte transmission is about 10 mS.

FIG. 15A illustrates the simplest, and lowest power form of the messaging protocol. A synchronization pulse 420 will be repeated every recurring period 410 (about one second in the exemplary embodiment). As a non-limiting example, the sync pulse on time may be about 40 mS. The recurring period 410 with at least a synchronization pulse 420 may be repeated indefinitely while the transmitter is on. Note that “synchronization pulse” is somewhat of a misnomer because the synchronization pulse 350 may be a steady frequency during the pulse period as illustrated by the “white” pulse 420′. The synchronization pulse 420 may also include signaling at the resonant frequency with the ON/OFF keying discussed above and as illustrated by the “hatched” pulse 420. FIG. 15A illustrates a minimal power state wherein power at the resonant frequency is supplied during the synchronization pulse 420 and the transmit antenna is off during a power period 450. All receive devices are allowed to receive power during the synchronization pulse 420.

FIG. 15B illustrates the recurring period 410 with a synchronization pulse 420, a reverse link period 430 and a power period 450′ wherein the transmit antenna is on and supplying full power by oscillating at the resonant frequency and not performing any signaling. The upper timing diagram illustrates the entire recurring period 410 and the lower timing diagram illustrates an exploded view of the synchronization pulse 420 and the reverse link period 430. The power period 450′ may be segmented into different periods for multiple receive devices as is explained below. FIG. 15B shows three power segments Pd1, Pd2, and Pdn for three different receive devices.

When forward link signaling occurs, the synchronization pulse 420 may include a warm-up period 422, a forward link period 424, and a listening period 426. The listening period 426 may include a handover period 427 and a beginning reverse link period 428. During the synchronization pulse 420, the transmitter may send out a forward link message during the forward link period 400 (indicated by the “hatched” section) and waits for a reply from a receiver during the listening period 426. In FIG. 15B, no receivers reply, which is indicated by the “white” sections during the listening period 426.

FIG. 15C is similar to FIG. 15B except that a receiver replies during the beginning reverse link period 428 and the reverse link period 430, as indicated by the “cross-hatched” sections. In FIG. 15, during the synchronization pulse 420, the transmitter sends out a forward link message during the forward link period 400 and waits for a reply from a receiver during the listening period 426. Any receivers that are going to reply begin their reply before the end of the handover period 427, during the beginning reverse link period 428, and possibly during the reverse link period 430.

As a non-limiting example, Table 2 shows some possible messages that may be sent by the transmitter and the receiver.

TABLE 2 TX Command TX message RX Reply RX message Null NDQ (New Device NDR (New DD TT PP rr cc Query) Device Response) DQ (Device Query) DD DS (Device DD TT PP cc Status) ACK (Acknowledge a device XX from previous DS) SA (Slot Assignment) DD NN MM cc RES (Reset all power slot assignments)

Where:

Null=no transmit command;

DD=Device number;

TT=Device Type;

PP=Power requested;

rr=a random number;

cc=a checksum;

NN=start of time slot; and

MM=end of time slot

In explaining table 1, the null command means that no messaging is sent by the transmitter during the forward link period 424. In line 2, a new device query (NDQ) is sent by the transmitter. If a receive device responds, it responds with a new device response (NDR) along with a device number (which should be zero for a new device, until the device number is assigned by the transmitter), a power request, a random number, and a checksum of all the data bits in the receive reply.

In line 3, a new device query (DQ) is sent by the transmitter along with a device number. The receive device that was addressed by the DQ replies with a device status (DS), along with the device number, the device type, the amount of power requested, and a checksum of all the data bits in the receive reply.

In line 4, the transmitter sends out an acknowledge (ACK) to the receiver that replied to the previous DQ. No receivers respond to an ACK

In line 5, the transmitter sends out a slot assignment (SA) along with a device number, a start time within the power period 450′, an end time within the power period 450′, and a checksum of all the data bits in the receive reply. No receivers respond to an SA.

In line 6, the transmitter sends out a reset (RES) indicating that all receivers should stop using their allocated time slots. No receivers respond to an RES. Of course, those of ordinary skill in the art will recognize that the commands and responses are exemplary and various embodiments contemplated within the scope of the present invention may use variations of these commands and responses, and additional commands and responses may be devised within the scope of the present invention.

To further illustrate how communication occurs, five different scenarios will be discussed. In the first scenario, initially no receive devices are within the coupling-mode region of the transmitter and one receive device enters the coupling-mode region. When no device are present in the coupling-mode region the transmitter will remain in the low power state as illustrated in FIG. 15A and repeat the synchronization pulse 420 every recurring period 410. The synchronization pulse 420 will include a NDQ during the forward link period 424 and the transmitter will listen for a reply during the listening period 426. If no reply is received, the transmitter shuts down until time for the synchronization pulse 420 of the next recurring period 410.

When a new receive device is introduced to the coupling-mode region, the receive device is initially on and listening for a synchronization pulse 420. The new receive device may use the synchronization pulse 420 for power but should go into a cloaked or non-power reception mode (referred to herein as “getting off the bus”) during the power period 450′. In addition, the new receive device listens for transmit commands and ignores all transmit commands except an NDQ. When a new receive device receive an NDQ, it remains on during the handover period 427, the beginning reverse link period 428, and possibly the reverse link period 430. After the forward link period 424 and before the end of the handover period 427, the receive device responds with a NDR, a device ID of zero (a new device ID will be assigned by the transmitter), a power amount request, a random number and a checksum. The new receive device then gets off the bus during the power period 450′.

If the transmitter receives the NDR correctly, it responds on the next synchronization pulse 420 with a slot assignment (SA) for the new receive device. The SA includes a device ID for the new receive device, a start time, an end time, and a checksum. The start time and end time for this SA will be zero indicating that the new receive device should not get on the bus for any time period during the power period 450′. The new receive device will receive a subsequent SA with actual start times and end times assigning a specific power segment Pdn when it can get on the bus. If the new receive device does not receive a proper checksum, in remains in new device mode and responds again to an NDQ.

In the second scenario, no receive devices are within the coupling-mode region of the transmitter and more than one receive device enters the coupling-mode region. In this mode, when two new receive devices are introduced to the coupling-mode region they are initially on the bus all the time. The new receive devices may use the synchronization pulse 420 for power but should get off the bus during the power period 450′ once a synchronization pulse 420 has been received. In addition, the new receive devices listen for transmit commands and ignore all transmit commands except an NDQ. When the new receive device receive an NDQ, they remain on during the handover period 427, the beginning reverse link period 428, and possibly the reverse link period 430. After the forward link period 424 and before the end of the handover period 427, the receive devices responds with a NDR, a device ID of zero (a new device ID will be assigned by the transmitter), a power amount request, a random number and a checksum.

However, since two or more receive devices are responding at the same time, and likely have different random numbers and checksums, the message received by the transmitter will be garbled, and the checksum in the transmitter will not be accurate. As a result, the transmitter will not send out a SA on the subsequent synchronization pulse 420.

When an immediate SA is not forthcoming after an NDR, each of the receive devices waits a random number of subsequent NDQs before responding with an NDR. For example, two devices both respond to the first NDQ so no subsequent SA happens. Device 1 decides to wait four NDQs before responding to another NDQ. Device 2 decides to wait two NDQs before responding to another NDQ. As a result, on the next NDQ sent out by the transmitter, neither device responds with an NDR. On the next NDQ sent out by the transmitter, only device 2 responds with an NDR, the transmitter successfully receives the NDR and sends out an SA for device 2. On the next NDQ, device 2 does not respond because it is no longer a new device and device 1 does not respond because its random waiting period has not elapsed. On the next NDQ sent out by the transmitter, only device 1 responds with an NDR, the transmitter successfully receives the NDR and sends out an SA for device 1.

In the third scenario, at least one receive device is in the coupling-mode region and a new receive device enters the coupling-mode region. In this mode, the new receive devices is introduced to the coupling-mode region and is initially on the bus all the time. The new receive devices may use the synchronization pulse 420 for power but should get off the bus during the power period 450′ once a synchronization pulse 420 has been received. In addition, the new receive devices listen for transmit commands and ignore all transmit commands except an NDQ. Periodically, the transmitter will issue an NDQ to see if any new devices have entered the coupling-mode region. The new device will then reply with an NDR. On the subsequent synchronization pulse 420, the transmitter will issue an SA for the new device with no power slots assigned. The transmitter then recalculates power allocation for all the devices in the coupling-mode region and generates new SAs for each device so there are no overlapping power segments Pdn. After each device receives its new SA, it begins getting on the bus only during its new Pdn.

In the fourth scenario, normal power delivery operation continues with no receive device entering or leaving the coupling-mode region. During this scenario, the transmitter will periodically ping each device with a device query (DQ). The queried device responds with a device status (DS). If the DS indicates a different power request, the transmitter may reallocate power allocation to each of the devices in the coupling-mode region. The transmitter will also periodically issues an NDQ as was explained above for the third scenario.

In the fifth scenario, a device is removed from the coupling-mode region. This “removed” state may be that the device is physically removed from the coupling-mode region, the device is shut off, or the device cloaks itself, perhaps because it does not need any more power. As stated earlier, the transmitter periodically sends out a DQ for all the devices in the coupling-mode region. If two consecutive DQs to a specific device do not return a valid DS, the transmitter removes the device from its list of allocated devices and reallocates the power period 450′ to the remaining devices. The transmitter will also assign the missing device a power allocation of zero time in case it is still receiving by is unable to transmit. If a device was erroneously removed from the power allocation, it may regain power allocation by responding to and NDQ with a proper NDR.

Table 3 illustrates a non-limiting sequence of commands and replies to illustrate how the communication protocol operates.

TABLE 3 Command Description Reply Description Comments DQ1 Query Device 1 DS 1 1 FF cc Device 1 is Cellphone with low type 1, wants battery max power DQ2 Query Device 2 DS 2 1 84 cc Device 2 is PDA with almost type 3, wants to charged battery reduce power time slot SA 2 84 Slot assign Reduce device 2's FF device 2 power slot (reduce first, then increase) SA 1 00 83 Slot assign Increase device 1's device 1 power slot NDQ New device NDR 00 04 FF rr cc New device Mouse with a low query found battery, max power SA 3 00 00 Slot assign Immediate reply after device 3 NDQ means it is for new device. Device ID is 3. Initial power slot is 0. SA 1 00 40 Slot assign Device 1 reassigned device 1 to ¼ power. SA 2 41 80 Slot assign Device 2 reassigned device 2 to ¼ power. SA 3 81 Slot assign Device 3 reassigned FF device 2 to ½ power. NDQ New device No reply so no new query device found. null DQ1 DQ2 DQ3 NDQ

Note that the first slot assignment for the new device allocates no time slot. Each existing device is allocated a new non-overlapping time slot, then the new device is finally allocated a time slot for receiving power.

In an exemplary embodiment, a wireless charging devices may display a visible signal, such as, for example, a light to the user indicating that it has successfully entered the charging region and registered itself to the local transmitter. This will give the user positive feedback that a device is indeed prepared to charge.

In other exemplary embodiments of the present invention, the receiver and transmitter may communicate on a separate communication channel 119 (e.g., Bluetooth, zigbee, cellular, etc) as is shown in FIG. 2. With a separate communication channel, the recurring period need not include any communication periods and the entire time may be devoted to the power period 450′. The transmitter may still allocate time slots to each receive device (communicated over the separate communication channel) and each receive device only gets on the bus for its allocated power segment Pdn.

The time-multiplexed power allocations described above may be the most-efficient method for supplying power to multiple receive devices within a transmitter's coupling-mode region. However, other power allocation scenarios may be employed with other embodiments of the present invention.

FIGS. 16A-16D are simplified block diagrams illustrating a beacon power mode for transmitting power between a transmitter and a one or more receivers. FIG. 16A illustrates a transmitter 520 having a low power “beacon” signal 525 when there are no receive devices in the beacon coupling-mode region 510. The beacon signal 525 may be, as a non-limiting example, such as in the range of 10 to 20 mW RF. This signal may be adequate to provide initial power to a device to be charged when it is placed in the coupling-mode region.

FIG. 16B illustrates a receive device 530 placed within the beacon coupling-mode region 510 of the transmitter 520 transmitting the beacon signal 525. If the receive device 530 is on and develops a coupling with the transmitter it will generate a reverse link coupling 535, which is really just the receiver accepting power from the beacon signal 525. This additional power, may be sensed by the load sensing circuit 216 (FIG. 12) of the transmitter. As a result, the transmitter may go into a high power mode.

FIG. 16C illustrates the transmitter 520 generating a high power signal 525′ resulting in a high power coupling-mode region 510′. As long as the receive device 530 is accepting power and, as a result, generating the reverse link coupling 535, the transmitter will remain in the high power state. While only one receive device 530 is illustrated, multiple receive devices 530 may be present in the coupling-mode region 510. If there are multiple receive device 530 they will share the amount of power transmitted by the transmitter based on how well each receive device 530 is coupled. For example, the coupling efficiency may be different for each receive device 530 depending on where the device is placed within the coupling-mode region 510 as was explained above with reference to FIGS. 8 and 9.

FIG. 16D illustrates the transmitter 520 generating the beacon signal 525 even when a receive device 530 is in the beacon coupling-mode region 510. This state may occur when the receive device 530 is shut off, or the device cloaks itself, perhaps because it does not need any more power.

As with the time-multiplexing mode, the receiver and transmitter may communicate on a separate communication channel (e.g., Bluetooth, zigbee, etc). With a separate communication channel, the transmitter may determine when to switch between beacon mode and high power mode, or create multiple power levels, based on the number of receive devices in the coupling-mode region 510 and their respective power requirements.

Exemplary embodiments of the invention include enhancing the coupling between a relatively large transmit antenna and a small receive antenna in the near field power transfer between two antennas through introduction of additional antennas into the system of coupled antennas that will act as repeaters and will enhance the flow of power from the transmitting antenna toward the receiving antenna.

In an exemplary embodiment, one or more extra antennas are used that couple to the transmit antenna and receive antenna in the system. These extra antennas comprise repeater antennas, such as active or passive antennas. A passive antenna may include simply the antenna loop and a capacitive element for tuning a resonant frequency of the antenna. An active element may include, in addition to the antenna loop and one or more tuning capacitors, an amplifier for increasing the strength of a repeated near field radiation.

The combination of the transmit antenna and the repeater antennas in the power transfer system may be optimized such that coupling of power to very small receive antennas is enhanced based on factors such as termination loads, tuning components, resonant frequencies, and placement of the repeater antennas relative to the transmit antenna.

A single transmit antenna exhibits a finite near field coupling mode region. Accordingly, a user of a device charging through a receiver in the transmit antenna's near field coupling mode region may require a considerable user access space that would be prohibitive or at least inconvenient. Furthermore, the coupling mode region may diminish quickly as a receive antenna moves away from the transmit antenna.

A repeater antenna may refocus and reshape a coupling mode region from a transmit antenna to create a second coupling mode region around the repeater antenna, which may be better suited for coupling energy to a receive antenna.

FIG. 17A illustrates a large transmit antenna 710A with a smaller repeater antenna 720A disposed coplanar with, and within a perimeter of, the transmit antenna 710A. The transmit antenna 710A and repeater antenna 720A are both formed on a table 740. A device including a receive antenna 730A is placed within the perimeter of the repeater antenna 720A. With very large antennas, there may be areas of the coupling mode region that are relatively week near the center of the transmit antenna 710A. Presence of this weak region may be particularly noticeable when attempting to couple to a very small receive antenna 730A. The repeater antenna 720A placed coplanar with the transmit antenna 710A, but with a smaller size, may be able to refocus the coupling mode region generated by the transmit antenna 710A into a smaller and stronger repeated coupling mode region around the repeater antenna 720A. As a result, a relatively strong repeated near field radiation is available for the receive antenna 730A.

FIG. 17B illustrates a transmit antenna 710B with a larger repeater antenna 720B with a coaxial placement relative to the transmit antenna 710B. A device including a receive antenna 730B is placed within the perimeter of the repeater antenna 720B. The transmit antenna 710B is formed around the lower edge circumference of a lamp shade 742, while the repeater antenna 720B is disposed on a table 740. Recall that with coaxial placements, the near field radiation may diminish relatively quickly relative to distance away from the plane of an antenna. As a result, the small receive antenna 730B placed in a coaxial placement relative to the transmit antenna 710B may be in a weak coupling mode region. However, the large repeater antenna 720B placed coaxially with the transmit antenna 710B may be able to reshape the coupled mode region of the transmit antenna 710B to another coupled mode region in a different place around the repeater antenna 720B. As a result, a relatively strong repeated near field radiation is available for the receive antenna 730B placed coplanar with the repeater antenna 720B.

FIG. 18A illustrates a large transmit antenna 710C with three smaller repeater antennas 720C disposed coplanar with, and within a perimeter of, the transmit antenna 710C. The transmit antenna 710C and repeater antennas 720C are formed on a table 740. Various devices including receive antennas 730C are placed at various locations within the transmit antenna 710C and repeater antennas 720C. As with the embodiment illustrated in FIG. 17A, the embodiment of FIG. 18A may be able to refocus the coupling mode region generated by the transmit antenna 710C into smaller and stronger repeated coupling mode regions around each of the repeater antennas 720C. As a result, a relatively strong repeated near field radiation is available for the receive antennas 730C. Some of the receive antennas are placed outside of any repeater antennas 720C. Recall that the coupled mode region may extend somewhat outside the perimeter of an antenna. Therefore, receive antennas 730C may be able to receive power from the near field radiation of the transmit antenna 710C as well as any nearby repeater antennas 720C. As a result, receive antennas placed outside of any repeater antennas 720C, may be still be able to receive power from the near field radiation of the transmit antenna 710C as well as any nearby repeater antennas 720C.

FIG. 18B illustrates a large transmit antenna 710D with smaller repeater antennas 720D with offset coaxial placements and offset coplanar placements relative to the transmit antenna 710D. A device including a receive antenna 730D is placed within the perimeter of one of the repeater antennas 720D. As a non-limiting example, the transmit antenna 710D may be disposed on a ceiling 746, while the repeater antennas 720D may be disposed on a table 740. As with the embodiment of FIG. 17B, the repeater antennas 720D in an offset coaxial placement may be able to reshape and enhance the near field radiation from the transmitter antenna 710D to repeated near field radiation around the repeater antennas 720D. As a result, a relatively strong repeated near field radiation is available for the receive antenna 730D placed coplanar with the repeater antennas 720D.

While the various transmit antennas and repeater antennas have been shown in general on surfaces, these antennas may also be disposed under surfaces (e.g., under a table, under a floor, behind a wall, or behind a ceiling), or within surfaces (e.g., a table top, a wall, a floor, or a ceiling).

FIG. 19 shows simulation results indicating coupling strength between a transmit antenna, a repeater antenna and a receive antenna. The transmit antenna, the repeater antenna, and the receive antenna are tuned to have a resonant frequency of about 13.56 MHz.

Curve 810 illustrates a measure for the amount of power transmitted from the transmit antenna out of the total power fed to the transmit antenna at various frequencies. Similarly, curve 820 illustrates a measure for the amount of power received by the receive antenna through the repeater antenna out of the total power available in the vicinity of its terminals at various frequencies. Finally, Curve 830 illustrates the amount of power actually coupled between the transmit antenna, through the repeater antenna and into the receive antenna at various frequencies.

At the peak of curve 830, corresponding to about 13.56 MHz, it can be seen that a large amount of the power sent from the transmitter is available at the receiver, indicating a high degree of coupling between the combination of the transmit antenna, the repeater antenna and the receive antenna.

FIG. 20A show simulation results indicating coupling strength between a transmit antenna and receive antenna disposed in a coaxial placement relative to the transmit antenna with no repeater antennas. The transmit antenna and the receive antenna are tuned to have a resonant frequency of about 10 MHz. The transmit antenna in this simulation is about 1.3 meters on a side and the receive antenna is a multi-loop antenna at about 30 mm on a side. The receive antenna is placed at about 2 meters away from the plane of the transmit antenna. Curve 810A illustrates a measure for the amount of power transmitted from the transmit antenna out of the total power fed to its terminals at various frequencies. Similarly, curve 840 illustrates a measure of the amount of power received by the receive antenna out of the total power available in the vicinity of its terminals at various frequencies. Finally, Curve 830A illustrates the amount of power actually coupled between the transmit antenna and the receive antenna at various frequencies.

FIG. 20B show simulation results indicating coupling strength between the transmit and receive antennas of FIG. 20A when a repeater antenna is included in the system. The transmit antenna and receive antenna are the same size and placement as in FIG. 20A. The repeater antenna is about 28 cm on a side and placed coplanar with the receive antenna (i.e., about 0.1 meters away from the plane of the transmit antenna). In FIG. 20B, Curve 810B illustrates a measure of the amount of power transmitted from the transmit antenna out of the total power fed to its terminals at various frequencies. Curve 820B illustrates the amount of power received by the receive antenna through the repeater antenna out of the total power available in the vicinity of its terminals at various frequencies. Finally, Curve 830B illustrates the amount of power actually coupled between the transmit antenna, through the repeater antenna and into the receive antenna at various frequencies.

When comparing the coupled power (830A and 830B) from FIGS. 20A and 20B it can be seen that without a repeater antenna the coupled power 830A peaks at about −36 dB. Whereas, with a repeater antenna the coupled power 830B peaks at about −5 dB. Thus, near the resonant frequency, there is a significant increase in the amount of power available to the receive antenna due to the inclusion of a repeater antenna.

As stated earlier, the receive antennas and transmit antennas are designed to operate in the near field of each other where they are tightly coupled to each other and their local environment. This means coupling and scattering of RF energy to and from other structures (which can include other devices that need charging) can influence the driving point impedance of both the transmit and receive antennas. A poor impedance match between the transmitter electronics and/or receive electronics to their respective antenna ports can cause degradation to the level of power transfer between the two. Further, since the local environment can change with time (devices and structures nearby being removed or added), the degradation can also be time dependant. The exemplary embodiments of the invention have the capability to adaptively improve the impedance match in response to this changing environment, thereby improving the power transfer between devices and the overall time to charge a device.

Exemplary embodiments of the invention include an RF circuit topology capable of adaptively optimizing the impedance match between the transmit electronics and transmit antenna or the receiver electronics and the receiver antenna. These adaptations are performed in a manner that self-adjusts to the local environment to optimize the power coupling between transmit antennas and receiver antennas.

Exemplary embodiments of the invention use feedback mechanisms and variable capacitors connected to the antenna to achieve this adaptive tuning function. As non-limiting examples, the adaptive tuning may be used to optimize a coupling of the near field radiation between a transmit antenna and a receive antenna, a transmit antenna and a repeater antenna, a repeater antenna and a receive antenna, or combinations thereof. In addition, the adaptive tuning function may be used to tune transmit antennas, receive antennas, or combinations thereof when additional receive antennas are placed in the coupling mode region of the transmit antenna.

FIG. 21A-21C are simplified block diagrams of adaptive tuning circuits for an antenna using a T-network, an L-network, and a Pi-network, respectively. FIGS. 21A-C include an antenna 904 with characteristic impedance L1 to receive or transmit a radiated field 106. The antenna is connected to a coupler 910 for splitting the signal between a voltage standing wave ratio (VSWR) detector 920 and a variable capacitor network, which may be designated generically as 950. Specifically, in FIG. 21A the variable capacitor network 950T is configured in a T-network, in FIG. 21B the variable capacitor network 950L is configured in an L-network, and in FIG. 21C the variable capacitor network 950P is configured in a Pi-network. The VSWR detector 920 is connected to a controller 930, which controls each of the variable capacitors using control signals. An RF signal 955 couples to additional circuitry (not shown) depending on whether the antenna 904 is part of a receiver, transmitter, or repeater.

In operation, the RF signal from the coupler 910 feeds the VSWR detector 920. As a non-limiting example, the VSWR detector 920 may be a diode power detector circuit, which generates a voltage that is proportional to the magnitude of the reflected signal. A large reflected signal at the antenna may indicate that the antenna 904 is operating inefficiently. This proportional voltage is read by the controller 930. The controller includes software to analyze the characteristics of the L-C network comprising the characteristic impedance L1 of the antenna 904 coupled with the variable capacitor network 950. Based on the proportional voltage, the controller 930 uses the control signals to adjust capacitance values of the variable capacitance networks.

In the case of FIG. 21A, the control signals may adjust capacitance for variable capacitors VC1A, VC2A, and VC3A configured in a T-network. In the case of FIG. 21B, the control signals may adjust capacitance for variable capacitors VC1B and VC2B configured in an L-network. In the case of FIG. 21C, the control signals may adjust capacitance for variable capacitors VC1C, VC2C, and VC3C configured in a Pi-network.

This adjustment of the variable capacitors closes the feedback loop. The proportional voltage from the VSWR detector 920 adjusts based on the new reflection characteristics at the antenna 904, the controller 930 samples the new proportional voltage and modifies the control signals to adjust the variable capacitor network 905 again. The feedback loop continues to monitor and adjust to minimize reflected power at the antenna 904, which maximized the power delivered out of the antenna 904.

FIG. 22 is a simplified block diagram of an adaptive tuning circuit for a transmit antenna 904 based on power consumption at the transmit antenna 904. An oscillator 922 drives a power amplifier 924, which receives power from a PA sensor 940 via a power input Vds. The PA sensor 940 feeds a controller 930, which generates control signals to adjust the variable capacitor (VC1C, VC2C, and VC3C) in the variable capacitor network 950P.

Instead of monitoring the change in reflected energy at the transmit antenna, the embodiment of FIG. 22 detects changes in the power consumed from the near field radiation 106. As the power consumed changes, reflected energy at the antenna may change, which may be minimized by adjusting the variable capacitor network 950P. The feedback mechanism operates similarly as for the embodiments of FIGS. 21A-21C, except that consumed power is monitored to determine antenna inefficiencies rather than a VSWR. Of course, while not illustrated, those of ordinary skill in the art will recognize any of the variable capacitor networks illustrated in FIGS. 21A-21C may be used in the FIG. 22 embodiment. Furthermore, other suitable variable capacitor networks may be used with any of the embodiments of FIGS. 21A-21C and 22.

FIGS. 23A and 23B are simplified circuit diagrams illustrating exemplary embodiments of variable capacitor networks. In FIG. 23A, the variable capacitors are implemented as switched capacitors configured in an L-network 950L. Thus, the L(RX/TX) is the characteristic impedance of the antenna, C11, C12, . . . C1n form the variable capacitor VC1B, and C21, C22, . . . C2m form the variable capacitor VC2B. The Z(load) models any load on the network. A 1:n decoder controls the switches to the VC1B capacitors and a 1:m decoder controls the switches to the VC2B capacitors. Thus, a variable amount of capacitance may be selected. The capacitors may be configured to all have the same capacitance, they may be configured to have binary weights, or they may be configured with other suitable weights.

In FIG. 23B, the variable capacitors are implemented as varactors (i.e., reverse biased variable capacitance diodes) configured in a T-network 950T. Thus, the control signals 935 are analog signals used to adjust the capacitance of variable capacitors VC1A, VC2A, and VC3A by adjusting the bias to diodes V11, V12, V21, V22, V31, and V32. As the bias is adjusted, the apparent capacitance of the combined reverse diodes (i.e., V11-V12, V21-V22, and V31-V32) adjusts.

Other types of variable capacitors may be used to implement the variable capacitor networks. As a non-limiting example, variable capacitor Micro-Electro-Mechanical System (MEMS) variable capacitors have been recently proposed, which may be used as the variable capacitors for any of the variable capacitor networks. Furthermore, more than one type of variable capacitor may be used in any given variable capacitor network.

FIGS. 24A and 24B illustrates simulation results for near field coupled transmit and receive antennas before and after adaptive tuning, respectively. In FIG. 24A, curve 860A illustrates the amount of power not transmitted from the transmit antenna at various frequencies (i.e., return loss). Similarly, curve 870A illustrates the amount of power not received by the receive antenna at various frequencies (i.e., return loss). Finally, curve 880A illustrates the amount of power actually coupled between the transmit antenna and the receive antenna at various frequencies. As can be seen in FIG. 24A, the transmit antenna is poorly tuned as illustrated by the relatively minor dip in the curve at the resonant frequency of about 13.56 MHz.

In FIG. 24B, curve 860B illustrates the amount of power not transmitted from the transmit antenna at various frequencies. Similarly, curve 870B illustrates the amount of power not received by the receive antenna at various frequencies. Finally, curve 880B illustrates the amount of power actually coupled between the transmit antenna and the receive antenna at various frequencies. As can be seen, the transmit antenna is tuned much better as indicated by the significant dip at the resonant frequency of about 13.56 MHz. The coupled energy indicated by curve 880B is not significantly higher in FIG. 24B relative to the coupled energy 880A in FIG. 24A. However, this may be due to other factors, such as, for example, the presence of another receive antenna, the placement of the receive antenna relative to the transmit antenna, or an improperly tuned receive antenna.

Thus, in systems with adaptive tuning in both the receive antenna and the transmit antenna, both antennas will be continuously tuned to adapt to changes in the environment, and even tuning changes at the other antenna. For example, in FIG. 24AAA curve 860A indicates that the receive antenna is not as well tuned as in FIG. 24B illustrated by curve 860B. This, change in tuning may be a result of the change in tuning in the transmit antenna. As a result, a receive antenna with adaptive tuning would attempt to achieve better tuning, resulting in more coupling between the transmit and receive antenna.

As stated earlier, an adaptive tuner network similar to those used with the receive and transit antennas can be used to further enhance the power coupling between transmit, receive, and repeater antennas. In the case of the repeater, RF-DC conversion circuitry (as illustrated by elements 308 and 310 in FIG. 11) may be integrated in the repeater antenna structure and can be used to convert RF power to DC power. This DC power may be used as the power source to control the repeater adaptive tuning network as well as other active elements in an active repeater. In this way, there may be no tethered power sources to an active repeater antenna. In an exemplary embodiment, a series of 3-4 LEDs may be integrated on the repeater and if practical, on the receiver antennas to indicate when each is in the range of the transmitter antenna.

Those of skill in the art would understand that information and signals may be represented using any of a variety of different technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips that may be referenced throughout the above description may be represented by voltages, currents, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination thereof.

Those of skill would further appreciate that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the embodiments disclosed herein may be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative components, blocks, modules, circuits, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the exemplary embodiments of the invention.

The various illustrative logical blocks, modules, and circuits described in connection with the embodiments disclosed herein may be implemented or performed with a general purpose processor, a Digital Signal Processor (DSP), an Application Specific Integrated Circuit (ASIC), a Field Programmable Gate Array (FPGA) or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.

The steps of a method or algorithm described in connection with the embodiments disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. A software module may reside in Random Access Memory (RAM), flash memory, Read Only Memory (ROM), Electrically Programmable ROM (EPROM), Electrically Erasable Programmable ROM (EEPROM), registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art. An exemplary storage medium is coupled to the processor such that the processor can read information from, and write information to, the storage medium. In the alternative, the storage medium may be integral to the processor. The processor and the storage medium may reside in an ASIC. The ASIC may reside in a user terminal. In the alternative, the processor and the storage medium may reside as discrete components in a user terminal.

In one or more exemplary embodiments, the functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented in software, the functions may be stored on or transmitted over as one or more instructions or code on a computer-readable medium. Computer-readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one place to another. A storage media may be any available media that can be accessed by a computer. By way of example, and not limitation, such computer-readable media can comprise RAM, ROM, EEPROM, CD-ROM or other optical disk storage, magnetic disk storage or other magnetic storage devices, or any other medium that can be used to carry or store desired program code in the form of instructions or data structures and that can be accessed by a computer. Also, any connection is properly termed a computer-readable medium. For example, if the software is transmitted from a website, server, or other remote source using a coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or wireless technologies such as infrared, radio, and microwave, then the coaxial cable, fiber optic cable, twisted pair, DSL, or wireless technologies such as infrared, radio, and microwave are included in the definition of medium. Disk and disc, as used herein, includes compact disc (CD), laser disc, optical disc, digital versatile disc (DVD), floppy disk and blu-ray disc where disks usually reproduce data magnetically, while discs reproduce data optically with lasers. Combinations of the above should also be included within the scope of computer-readable media.

The previous description of the disclosed exemplary embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these exemplary embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein. 

1. A wireless power receiver, comprising: a receive antenna for coupling with a transmit antenna through a near field radiation generated by the transmit antenna to generate an RF signal; a voltage standing wave ratio detector operably coupled to the receive antenna to generate a mismatch signal indicating a degree of mismatch at the receive antenna; a controller operably coupled to the mismatch signal to analyze the mismatch signal and generate control signals responsive thereto; and a variable capacitor network operably coupled to the control signals and between the receive antenna and the RF signal, the variable capacitor network to modify a resonance characteristic of the receive antenna and the variable capacitor network by modifying a capacitance of the variable capacitor network responsive to the control signals.
 2. The wireless power receiver of claim 1, wherein the variable capacitor network comprises a network selected from the group consisting of a T-network, an L-network, and a Pi-network.
 3. The wireless power receiver of claim 1, wherein the variable capacitor network comprises a plurality of variable capacitors, each variable capacitor controlled by a control signal of the control signals, wherein a capacitance of each capacitor is adjusted responsive to its corresponding control signal.
 4. The wireless power receiver of claim 3, wherein each variable capacitor of the plurality comprises a plurality of switched capacitors operably coupled in parallel, each switched capacitor comprising a switch and a capacitor operably coupled in series, wherein each switched capacitor is selectively included in the variable capacitor responsive to the control signal operatively coupled thereto.
 5. The wireless power receiver of claim 3, wherein each variable capacitor of the plurality comprises at least one varactor operatively coupled to a corresponding control signal acting as a bias control for the at least one varactor, the at least one varactor selected from the group consisting of a reverse bias varactor diode and a MEMS variable capacitor.
 6. A wireless power transmitter, comprising: a transmit antenna for generating a near field radiation for coupling to a receive antenna; a voltage standing wave ratio detector operably coupled to the transmit antenna to generate a mismatch signal indicating a degree of mismatch at the transmit antenna; a controller operably coupled to the mismatch signal to analyze the mismatch signal and generate control signals responsive thereto; and a variable capacitor network operably coupled to the control signals and between the transmit antenna and an RF signal, the variable capacitor network to modify a resonance characteristic of the transmit antenna and the variable capacitor network by modifying a capacitance of the variable capacitor network responsive to the control signals.
 7. The wireless power transmitter of claim 6, wherein the variable capacitor network comprises a network selected from the group consisting of a T-network, an L-network, and a Pi-network.
 8. The wireless power transmitter of claim 6, wherein the variable capacitor network comprises a plurality of variable capacitors, each variable capacitor controlled by a control signal of the control signals, wherein a capacitance of each capacitor is adjusted responsive to its corresponding control signal.
 9. The wireless power transmitter of claim 8, wherein each variable capacitor of the plurality comprises a plurality of switched capacitors operably coupled in parallel, each switched capacitor comprising a switch and a capacitor operably coupled in series, wherein each switched capacitor is selectively included in the variable capacitor responsive to the control signal operatively coupled thereto.
 10. The wireless power transmitter of claim 8, wherein each variable capacitor of the plurality comprises at least one varactor comprising a reverse bias diode operatively coupled to a corresponding control signal acting as a bias control for the at least one varactor.
 11. A wireless power transmitter, comprising: a transmit antenna for generating a near field radiation for coupling to a receive antenna; an amplifier with a power signal for supplying power to the amplifier, the amplifier for applying an RF signal to the transmit antenna; a load sensing circuit operably coupled to the amplifier and for detecting a change in current through the power signal, wherein the load sensing circuit generates a load indication signal responsive to the change in current; a controller operably coupled to the load indication signal and to generate control signals responsive to the load indication signal; and a variable capacitor network operably coupled to the control signals and between the transmit antenna and the RF signal, the variable capacitor network to modify a resonance characteristic of the transmit antenna and the variable capacitor network by modifying a capacitance of the variable capacitor network responsive to the load indication signal.
 12. The wireless power transmitter of claim 11, wherein the variable capacitor network comprises a network selected from the group consisting of a T-network, an L-network, and a Pi-network.
 13. The wireless power transmitter of claim 11, wherein the variable capacitor network comprises a plurality of variable capacitors, each variable capacitor controlled by a control signal of the control signals, wherein a capacitance of each capacitor is adjusted responsive to its corresponding control signal.
 14. The wireless power transmitter of claim 13, wherein each variable capacitor of the plurality comprises a plurality of switched capacitors operably coupled in parallel, each switched capacitor comprising a switch and a capacitor operably coupled in series, wherein each switched capacitor is selectively included in the variable capacitor responsive to the control signal operatively coupled thereto.
 15. The wireless power transmitter of claim 13, wherein each variable capacitor of the plurality comprises at least one varactor comprising a reverse bias diode operatively coupled to a corresponding control signal acting as a bias control for the at least one varactor.
 16. A method, comprising: generating an electromagnetic field at a resonant frequency of a transmit antenna to create a coupling-mode region within a near field of the transmit antenna; disposing a receive antenna within the coupling-mode region, wherein the receive antenna resonates substantially near the resonant frequency; and adaptively tuning a tunable antenna selected from the group consisting of the transmit antenna and the receive antenna by: detecting a mismatch at the tunable antenna by generating a mismatch signal responsive to a voltage standing wave ratio at the tunable antenna; and modifying a resonance characteristic of the tunable antenna by adjusting a capacitance of a variable capacitor network operably coupled to the tunable antenna.
 17. The method of claim 16, wherein the transmit antenna comprises a first tunable antenna and the receive antenna comprises a second tunable antenna and the acts of detecting the mismatch and modifying the resonance characteristic are applied to the transmit antenna and the receive antenna.
 18. The method of claim 16, wherein adjusting the capacitance of the variable capacitor network comprises selectively enabling a plurality of switched capacitors responsive to the mismatch signal, wherein the plurality of switched capacitors are operably coupled to form the variable capacitor network.
 19. The method of claim 16, wherein adjusting the capacitance of the variable capacitor network comprises biasing a plurality of varactors responsive to the mismatch signal, wherein the plurality of varactors are operably coupled to form the variable capacitor network.
 20. A wireless power transfer system, comprising: means for generating an electromagnetic field at a resonant frequency of a transmit antenna to create a coupling-mode region within a near field of the transmit antenna; means for receiving the resonant frequency within the coupling-mode region with a receive antenna, wherein the receive antenna resonates substantially near the resonant frequency; and means for adaptively tuning a tunable antenna selected from the group consisting of the transmit antenna and the receive antenna using: means for detecting a mismatch at the tunable antenna by generating a mismatch signal responsive to a voltage standing wave ratio at the tunable antenna; and means for modifying a resonance characteristic of the tunable antenna by adjusting the capacitance of a variable capacitor network operably coupled to the tunable antenna.
 21. The system of claim 20, wherein: the transmit antenna comprises a first tunable antenna with the means for detecting the mismatch and the means for modifying the resonance characteristic; and the receive antenna comprises a second tunable antenna with the means for detecting the mismatch and the means for modifying the resonance characteristic.
 22. The system of claim 20, wherein the adjusting the capacitance of the variable capacitor network comprises selectively enabling a plurality of switched capacitors responsive to the mismatch signal, wherein the plurality of switched capacitors are operably coupled to form the variable capacitor network.
 23. The system of claim 20, wherein adjusting the capacitance of the variable capacitor network comprises biasing a plurality of varactors responsive to the mismatch signal, wherein the plurality of varactors are operably coupled to form the variable capacitor network. 